It is desirable in many applications to impress a current pulse having the steepest edges possible of a predetermined current amplitude on a waveguide. This is, for example, useful in magnetostrictive position measurements. In this method of position measurements, a tangentially pre-magnetized wire or a tangentially pre-magnetized pipe is mechanically biased via magnetostriction by an axially magnetized position sensor. By an electrical current pulse sent through the wire or through the pipe, a soundwave propagating along the wire or pipe is triggered at the location of the magnetic position sensor. This mechanical wave deflects the tangential elementary magnets at the location of the wavefront, resulting in a magnetic wave along the wire and/or pipe propagating together with the soundwave. When arriving at the end of the assembly, the magnetic wave is detected using a coil. The time between the initial current impressed on the wire and/or pipe and detecting the magnetic wave in the coil is a very precise measure of the position of the sensor magnet which in known systems can be determined to a position of a few μm.
One of the challenges in magnetostrictive position measurements is impressing a steep-edge current impulse on the waveguide used, i.e. on the tangentially pre-magnetized wire and/or the pre-magnetized pipe. In known solutions current sources realized discretely by means of MOS transistors, wherein the current sources could be switched on and off have been used for impressing the current. The supply voltage in conventional circuit assemblies of this kind usually is between 5 volts and 40 volts. How fast the current reaches its final value is, in assemblies of this kind, not only dependent on how fast the gate of the switched transistor reaches the “on” voltage, but also on the transient behavior of the measuring distance. The measuring distance for fast processes is, above all, defined by its waveguide characteristics. Due to the very often high electrical characteristic impedance of the conductor or conductors used, the current, with small supply voltages, takes very long until reaching the final value of about 1 to 3 amperes. With a supply voltage of 5 volts and a wave resistance of the conductor used of, for example, 100 ohms, when ideally fast switching on the transistor, in the first moment after switching on, only a current of 5 V/100 ohms=50 mA can be impressed on the conductor. Then, the electrical wave at first has to propagate along the conductor. At the end of the line and/or the conductor, there is a short circuit allowing the high current flow. The wave is reflected there and after the return of the wave, another 50 mA can be impressed up to the beginning of the conductor. Consequently, the current increases over time in dependence on the length of the line and the supply voltage available.
The corresponding behavior matches the observation that a short-circuited line can be considered to be an inductance for not-too-high frequencies where the electrical length of the line is considerably shorter than a fourth of the wavelength. However, an inductance delays the increase in the current flow.
An improvement in the slew rate of the current pulse can be achieved by using higher a supply voltage allowing the full final amount of the current pulse to be achieved already in a first step. With a characteristic impedance of 220 ohms and a current final value of 1 ampere, a supply voltage of 220 volts is necessary. This voltage has to be generated in addition to the supply voltages present of the remaining electronics. This means considerably increased expenses for components increasing the area consumption and causing considerable additional cost.
FIG. 7 shows a circuit diagram of a circuit assembly for impressing a current pulse on a waveguide according to known solutions. The circuit assembly in its entirety is referred to by 800. The circuit assembly 800 includes a voltage source 810, a switching and regulating assembly 812, a waveguide 814 and a waveguide termination 816. The waveguide here includes a first conductor LTRA1 and a second conductor LTRA. The first conductor LTRA1 is coupled to the positive terminal of the voltage source 810 at the first end 820 of the waveguide 814. The second conductor LTRA of the waveguide 814 is coupled to the switching and regulating circuit 812 at the first end 820. The first conductor LTRA1 and the second conductor LTRA of the waveguide are connected to each other at the second end 830 of the waveguide 814 via a waveguide termination 816. The waveguide termination 816 here is formed by a resistor. The switching and regulating circuit 812 includes an n-channel MOS field-effect transistor 840 of the enhancement type the drain-source distance of which is connected in series to a resistor 842 between the terminal NIN of the second conductor LTRA of the waveguide 814 and the reference potential GND. A protection diode 844 protecting the transistor from great negative voltage peaks when switching off is connected in parallel to the drain-source distance of the n-channel MOS field-effect transistor. Furthermore, the gate terminal of the transistor is driven by a pulsed voltage source 846.
Based on the structural description of the circuit assembly 800 according to a known solution, the mode of functioning thereof will be discussed below. The starting state here is a currentless state of the waveguide 814, i.e. the switching and regulating circuit 812 has been switched off and/or has been in a high-impedance state for a sufficiently long period of time. If the switching and regulating circuit 812 is enabled by a suitable driving by the voltage source 846, i.e. if the n-channel MOS field-effect transistor 840 is placed in a conducting state, a voltage UIN which is in a context of UIN=ZW*IIN with the corresponding input current IIN will be applied to the first end 820 of the waveguide 814 via the terminals PIN and NIN. ZW here is the waveguide 814 characteristic impedance. The current IIN can be regulated by the switching and regulating circuit 812 in dependence on the gate potential of the n-channel MOS field-effect transistor 840 determined by the voltage source 846. If, for example, the voltage source 810 has a voltage of 250 volts and additionally the characteristic impedance of the waveguide 814 equals 220 ohms, the input voltage UIN of the waveguide at a current IIN of 1 ampere will be UIN=220 volts. Thus, a voltage of 30 V=250 V−220 V is across the switching and regulating circuit 812.
If the waveguide is, as described before, terminated in an impedance-correct manner by a waveguide termination 816, no signal reflection will occur at the second end 830 of the waveguide 814. The current flow through the waveguide 814 thus is approximately 1 ampere independent on time, until switching off the current flow, wherein it is assumed that the switching and regulating circuit 812 performs a stable regulation of the current flow.
The current flow can also be interrupted when suitably driving the n-channel MOS field-effect transistor 840 by the voltage source 846. Due to the termination of the transmission line 840 having the correct impedance, there are no signal reflections and the waveguide 814 will be currentless one signal runtime after switching off the n-channel MOS field-effect transistor 840.
The conventional known circuit assembly 800 has a number of disadvantages. For usual waveguide impedances (exemplarily 220 ohms) and technically usual currents (exemplarily 1 ampere), the voltage source 810 has to have very high a voltage (exemplarily 250 volts). Technically, this is very unfavorable since such a voltage is not directly available in conventional systems and has to be generated specially, entailing considerable cost for realization. In addition, conventional circuit assemblies are of disadvantage in that the second end 830 of the waveguide 814 is terminated by a resistor 816. In a quasi-stationary state, the same current passes through the terminating resistor as through the waveguide 814. This generates a considerable power loss in the terminating resistor 816. This is very difficult to dissipate and also decreases the efficiency of the circuit assembly highly so that the current consumption of an overall measuring assembly increases greatly.